The present invention is generally related to integrated circuits (ICs) and, more particularly, to measuring the quality of delay tests used to test ICs for faults, or defects.
After an IC is manufactured, it is tested on an IC tester before it is shipped to a customer. The goal of the testing is to verify that each individual IC was manufactured correctly without defects. At the highest level, testing may be reduced to simply plugging the IC into a host system and determining whether the system appears to be functioning normally while running applications. If the system works, then a determination may be made that the IC is ready to be shipped. However, this type of system-level test does not ensure that the IC is defect free, since the given applications used may exercise only a subset of the IC""s functionality. This type of high-level system-based testing also requires a relatively large amount of time.
An alternative to the system test approach is known as functional testing. This type of testing is performed on a general-purpose IC tester (known as Automated Test Equipment, or ATE). This type of testing applies logic 1s and/or 0s to the input pins of the IC in order to stimulate all of the logic gates within the IC, and determines whether each logic gate outputs the correct result by observing the output pins of the IC. The patterns applied to and the results expected from each IC pin are stored in memory on the ATE and exercise the various functional aspects of the IC. If the IC responds correctly to all test stimuli, it is considered to be of shipment quality. However, given the complexity and sequential depth of modem ICs, creating a sufficiently thorough test to be applied via the pins is very difficult, and given the large number of pins on some ICs, the cost of the ATE resources can become prohibitive.
A third alternative to the system testing and functional testing approaches is known as structural testing. Instead of exercising the functional aspects of the IC, this type of testing applies logic 1s and/or 0s internally to stimulate all of the logic gates within the IC, and determines whether each logic gate outputs the correct result, again internally. This internal controllability and observability is obtained by using modified memory elements (flip-flops) inside the IC that are serially connected into a scan chain during test mode. This well-known technique of xe2x80x9cscan designxe2x80x9d has been in wide use for many years. In xe2x80x9cfull scanxe2x80x9d designs, every internal flip-flop in the IC is made xe2x80x9cscannablexe2x80x9d by adding a serial access to a predecessor flip-flop and a successor flip-flop on the scan path during test mode. Thus, all the logic gates on the IC are surrounded by scannable flip-flops and become combinationally testable. In order to perform a scan test, data is serially shifted into all of the flip-flops in the scan path while the IC is in test mode, the resulting response of the logic gates to the final scanned-in state stimulus is captured by clocking all the flip-flops one or more times while the IC is in normal mode, and then by serially shifting the newly captured data out of the IC while in test mode. The captured data is analyzed by the ATE as it is shifted out to determine whether the correct results were obtained. The ATE is also responsible for switching the IC between normal and test modes appropriately as well as for providing the clock stimulus.
In order to create a structural test, a software tool uses a simulation model of the IC, which includes the scan flip-flops and all of the combinational logic of the IC. A xe2x80x9cfault modelxe2x80x9d that represents hypothesized defects is superimposed on the simulation model of the IC in order to guide the creation of specific test patterns (also called test vectors) that are intended to expose faulty logic gates. The software tool then generates test patterns for each location in the IC model at which a fault, or defect, could exist. Each test pattern is a set of 1s and 0s that are necessary to excite and propagate the hypothesized fault to an observation point (i.e. a scannable flip-flop), as well as the expected response of a defect-free IC. If an IC responds to such a pattern with data other than that expected, then the hypothesized fault is deduced to be present and the IC is thus determined to be defective and is not shipped. The complete set of test patterns (called a test sequence or a test set) is intended to cover all possible faults in an IC.
The spectrum of all possible faults in an IC is, unfortunately, very broad. While many defects result in permanent logical errors that can be easily detected by scan-based tests, some defects manifest themselves only as increased delays in the IC. Therefore, if the scan test is performed without taking the speed at which the gates should respond into account, such xe2x80x9cdelay defectsxe2x80x9d may go undetected. For example, assuming a NOR gate that has a weak pull down transistor, the gate may produce the correct logical value if given enough time, but will not produce the value correctly under the timing specifications for the IC. Therefore, each gate must be checked to determine whether its logical function is correctly performed and whether it is performed in a timely fashion. A pattern that does not take timing into account is called a xe2x80x9cstaticxe2x80x9d test, while one that does execute under timing constraints is called a xe2x80x9cdynamicxe2x80x9d test. A dynamic test for a given logic gate is created by running two test patterns in sequence at full clock speed and determining whether a slow-to-rise (STR) or slow-to-fall (STF) delay fault exists.
For example, if one input of a two-input NOR gate is held at 0 for two clock cycles while the other input changes from a 0 on the first clock cycle to a 1 on the second clock cycle, the output should change from a 1 on the first clock cycle to a 0 on the second clock cycle. If the output does not change from a 1 to a 0 within specified timing margins, a slow-to-fall fault exists. Similarly, if one input of the NOR gate is held at 0 for two clock cycles while the other input changes from a 1 on the first clock cycle to a 0 on the second clock cycle, the output should change from a 0 on the first clock cycle to a 1 on the second clock cycle. If the output does not change from a 0 to a 1 within specified timing margins, a slow-to-rise fault exists.
FIG. 1 is a block diagram illustrating a series of scan flip-flops 1, 2, and 3 as well as combinational logic 4 of an IC that incorporates the aforementioned scan design and that can be used for performing static and dynamic tests. Each of the flip-flops 1, 2, and 3 has its data input, D, connected to the output of one of the multiplexers 6, 7, or 8, respectively. When the scan enable signal, SC_EN is low (i.e., not asserted), the data at input 0 of the multiplexers 6, 7, and 8 is captured by the flip-flops 1, 2, and 3, respectively, on the rising edge of the clock, CLK. Therefore, when the scan enable signal SC_EN is not asserted, the IC is functioning in the normal operational manner. The combinational logic 4 will normally receive a plurality of primary input signals 9 originating from the input pins of the IC and will drive a plurality of primary output signals 11 that terminate at the output pins of the IC. The scan chain begins at input S_I 12 and ends at output S_O 13. The flip-flops act as a serial shift register between these two points when the scan enable signal SC_EN is asserted.
The black dots separating flip-flops 2 and 3 are intended to indicate that the scan chain may, and normally does, include many more flip-flops than the three shown in the figure (e.g., 100,000 flip-flops on a contemporary IC is not uncommon). It should be noted that corresponding additional inputs and outputs of the combinational logic 4 are connected to these additional scan flip-flops not shown in FIG. 1 to enable every gate of the combinational logic to be tested using the scan design technique. It should also be noted that several independent scan chains can be used to link groups of flip-flops together instead of one long scan chain as shown.
Given this scan design architecture, several varieties of tests are possible. The first is to establish that the scan chain itself is functional. This is accomplished by asserting the scan enable signal SC_EN (i.e., setting it to logic 1 in this example scan design configuration) and then cycling the clock as many times as there are flip-flops on the scan chain to load the chain, then again cycling the clock that many times to unload the scan chain. The signals at the scan in inputs, S_I, of the multiplexers 6, 7, and 8 are captured by the flip-flops 1, 2, and 3, respectively, on each clock cycle. The first flip-flop on the scan chain captures the S_I input pin 12, and the last flip-flop on the scan chain drives the S_O output pin 13. The scan logic is functioning as one large shift register during this test, with each flip-flop outputting a data value on the rising edge of the clock CLK. During the second half of this test, the expected response of the S_O pin 13 should match the values delivered on the S_I pin 12 in the first half of the test.
For example, assuming the scan design logic shown in FIG. 1 comprises 100 flip-flops, 200 clock cycles will be issued to shift a 100-bit test pattern of 1s and/or 0s into the flip-flops and completely out again; SC_EN will be held high over the 200 clock cycles. If all 100 bits appear properly on the S_O pin 13 in order during the second half of the test, then the scan chain is operational.
Once a determination is made that the flip-flops are functioning properly as a scan chain, the combinational logic 4 can be tested. In this case, again assuming 100 flip-flops in the scan design chain, the SC_EN signal will be asserted and 100 clock cycles will be issued to enable a 100-bit test pattern to be shifted into the flip-flops. The final state shifted in at this point in the test is available at the Q-outputs of the flip-flops and corresponds to the stimulus portion of this test pattern. After the combinational logic 4 has settled, the circuit response is available at the 0 inputs of the multiplexers and is captured in the flip-flops by pulsing the clock CLK exactly once with the SC_EN signal held low. Then, by holding the SC_EN signal high and issuing 100 clock cycles, the captured response data is shifted out for analysis. The data shifted out is analyzed by the ATE to determine if the circuit responded to the stimulus properly (against the stored expected response as a reference).
Whether or not this type of test is static or dynamic depends upon the relative timing of the clock signal CLK and the test mode signal SC_EN. A static test would result when there is a pause between the application of the last shift action (the 100th clock pulse with SC_EN high in this example) and the application of the capture event (the single clock pulse with SC_EN low). There are two well-known methods for applying dynamic tests. In the first, known as xe2x80x9clast-shift-launchxe2x80x9d or xe2x80x9cskewed-loadxe2x80x9d testing, the SC_EN signal must be capable of being switched from high to low in between two at-speed clock pulses (the 100th and 101st, in this example). This would apply two test patterns executed in sequence at speed, with the transitions caused by the last shifted state (from CLK 99 to CLK 100) in test mode being captured by CLK 101 in normal mode (just after SC_EN was lowered at speed). Although this is difficult to do, it has been accomplished in the past by carefully designing the SC_EN signal. Due to the difficulty of making the scan enable signal operate at speed, some designs perform delay testing at reduced speed, with consequent reduction in test quality. Alternatively, the second scheme for implementing dynamic tests, known as xe2x80x9cfunctional justificationxe2x80x9d, or xe2x80x9cbroadsidexe2x80x9d delay testing, relaxes this demand on the SC_EN signal, but requires two at-speed clocks during the time when SC_EN is low (i.e., during normal operation). This technique loops through the combinational logic 4 not once but twice, making the burden on the test pattern generation tools twice as great. For example, after the 100th pulse of the clock CLK with the scan enable signal SC_EN high, SC_EN can be lowered at leisure (at any desired time), then the clock CLK is pulsed twice in succession at full chip speed. The transitions launched on the first of these two clock pulses are captured by the second pulse. The scan enable signal SC_EN is the raised at leisure, then 100 clock pulses are applied to shift out the captured data. It should be noted that the combinational logic 4 is exercised twice during the time when SC_EN was low.
In whatever manner dynamic tests are implemented, they are intended to detect delay faults. There are two popular delay fault models: transition faults (also known as gate delay faults) and path delay faults. A transition fault models the situation where a single gate in an IC is slow (i.e., it will produce the correct logical output, but not in a timely fashion). A path delay fault models the situation where several gates are marginally slow, such that any one of them won""t adversely affect the ability of the circuit to run at speed, but the combination of all the incremental delays will cause the circuit to be too slow when a path connecting those gates is sensitized.
The quality of tests generated using the transition fault model is negatively affected by the xe2x80x9cgreedyxe2x80x9d nature of test generation algorithms, which tend to select the easiest (i.e., shortest) route into and out of the gate in question, even when there are other (longer) routes that would make the transition test more sensitive to a given delay defect. The measurement of transition test quality has traditionally been in terms of whether or not a given transition fault was detected at all; no accounting for the exact paths used to implement the transition tests is made.
The manner in which transition fault testing suffers quality reduction due to short path selection will be described with reference to the schematic diagram of FIG. 2. The schematic diagram of FIG. 2 shows a simple example of what might be found in the combinational logic 20 of an IC. The logic 20 comprises, among other logic gates, an AND gate 21, a NOR gate 22, and a NAND gate 23, which are connected such that they form a portion of a logical path connecting circuit nodes B, C, D, and E. Also shown in the figure are two scan flip-flops 24 and 25, as well as a delay fault location marked with an xe2x80x9cXxe2x80x9d at the output of NOR gate 22.
Present transition fault test generation algorithms often take advantage of the direct controllability and observability, respectively, afforded by the scan flip-flops 24 and 25 and launch and capture a rising transition passing through the fault site by using these short paths into and out of the fault site. This choice, unfortunately, reduces the quality of the transition test pattern, since the length of the path through the fault site is not taken into account in determining whether or not a transition fault is detected by a test pattern. For example, assuming a slow-to-rise (STR) fault is located at the xe2x80x9cXxe2x80x9d on the output of NOR gate 22, a test pattern which produces two consecutive 0s from AND gate 21 at node C on two consecutive clock cycles while launching a falling transition from flip-flop 24 will be able to observe the fault effect at node D via flip-flop 25 on the second clock cycle. Specifically, if the input to flip-flop 25 does not change from 1 to 0 within the second clock cycle, a determination is made that a transition fault exists. If, however, the input to flip-flop 25 does fall from 1 to 0 within the second clock cycle, a determination is made that no defect exists. The problem with this determination is that a delay defect at said fault site could indeed exist, but could be too short in duration to affect the particular (short) path Q-D taken to test it. If, alternatively, the fault were tested by holding the output Q of flip-flop 24 at a constant logic 0 while propagating a transition along path A-B-C-D-E-F, or some other relatively long path through the fault site, then the determination that a passing test does in fact indicate the absence of a defect can be made with much more confidence.
There are three important time intervals with respect to the timing of a particular path: 1) the clock period, T, of the system clock which is connected to the flip-flops on either end of the path; 2) the delay of the path, D, which measures the time interval between the launch of a transition at the start of the path until the arrival of the transition at the end of the path; and 3) the slack of the path, S, which measures the difference between the arrival time of the transition at the end of the path and the next rising edge of the system clock (which is simply expressed as S=Txe2x88x92D). Timing analysis tools typically express their results in terms of path slack. Therefore, the expression xe2x80x9cpath slackxe2x80x9d, or simply xe2x80x9cslackxe2x80x9d, will be used herein to denote this variable.
The impact of path selection on the quality of a delay fault test will be explained with reference to the timing diagrams in FIG. 3A and FIG. 3B. Both figures depict the timing waveforms for both a short path and a long path with respect to the clock waveform. FIG. 3A represents the defect-free case and FIG. 3B represents the case in which a delay defect is present. FIG. 3A shows, in order from top to bottom: the waveform 26 for the clock with a period of T time units (typically nanoseconds); the waveform 27 for a node representing a common launch point for both the short and the long paths, which experiences a transition one clock-to-Q delay time (T_ck_q) after the start of the clock period; the waveform 28 at the end of the short path, showing that the resultant transition occurs with plenty of slack time (SLACK_short_path) prior to the second rising edge of the clock; and, finally, the waveform 29 at the end of the long path, showing that its resultant transition occurs with very little slack time (SLACK_long_path) prior to the second rising edge of the clock.
It is clear that a delay defect on the long path could easily push the resultant transition beyond the second rising clock edge, which would case the delay test to fail since the wrong value would be captured. Exactly this situation is illustrated in FIG. 3B, which shows, in order from top to bottom: 1) the waveform 31 for the clock with a period of T time units (typically nanoseconds); 2) the waveform 32 for a node representing a common launch point for both the short and the long paths, which experiences a transition (shown in solid lines) delayed by a defect somewhat later than in the defect-free case (shown in dashed lines); 3) the waveform 33 at the end of the short path, showing that the resultant transition still occurs in time to beat the second rising clock edge (shown by the passing margin time); and, finally, 4) the waveform 34 at the end of the long path, showing that its resultant transition occurs well after the second rising clock edge (shown by the failing margin time). In this example, a test using the short path to test for the defect on the common launch point would falsely pass an IC manufactured with a delay defect of the duration shown, whereas a test using the long path would correctly identify the IC as defective. This is evident from the fact that the passing margin 35 associated with the short path 33 overlaps the failing margin 36 associated with the long path 34. It is clear that measuring the quality of transition fault tests must account for the actual path used to detect the faults.
While the path delay fault model obviously does not suffer from the problem of using too short a path through a fault site, since, by definition, the path delay fault model fully specifies the path in question as comprising the fault site, it is, along with the transition fault model, subject to quality reduction by another factor, namely the actual speed of application of the test. The timing diagram in FIG. 4 represents the use of three different clock periods: 1) nominal 41, 2) slow 42, and 3) fast 43, respectively, to apply a delay test to a path. The bottom waveform 44 indicates the substantial change in timing slack that occurs as the clock period is varied. A relatively short path can be tested much more stringently by simply decreasing the period of the system clock (i.e., increasing the frequency of the system clock) such that the excess slack time is reduced thereby increasing the sensitivity of the test to smaller delay defects. Conversely, if the clock period is increased (due to inadequate ATE resources that limit the frequency at which clocks are applied, etc.), all paths will acquire additional slack and all tests will become less sensitive to small delay defects, risking the passage of a defective IC that could later fail in its normal operating environment at fill clock speed.
Therefore, transition fault tests and path delay tests can both be rendered less effective by this technique of slowing down the system clock. It is also important to note that speeding up the system clock to improve test quality will work only to the point at which the slack of the longest paths tested becomes negative as the clock period is decreased. At this point, every IC will fail the test (because the test has been made too stringent, in violation of the maximum design specification for the clock operating frequency).
Currently, quality metrics such as the length of the specific path used to detect a delay fault and the application speed on the tester are not used to measure the quality of delay tests. Consequently, tools used to generate test patterns also do not take such quality metrics into account in generating tests. Accordingly, a need exists for a technique that measures the quality of delay tests in order to ensure that the tests are capable of determining whether a detected defect is a defect that may make the IC perform improperly during operation.
In accordance with the present invention, a method and apparatus are provided for determining quality metrics associated with a set of test patterns used to test an integrated circuit (IC) for delay defects. Metrics are formulated for the test of each individual delay fault, and these individual metrics are collected into respective IC-wide metrics. The key metrics, both at the individual fault level and the IC level, are referred to herein as the xe2x80x9cDelay Defect Exposurexe2x80x9d (or DDE, expressed in time units, such as nanoseconds), and the xe2x80x9cDelay Sensitivity Ratioxe2x80x9d (or DSR, expressed as a percentage). For both metrics, the delay time relative to the appropriate system clock associated with a longest sensitizable path through the IC that includes the fault site is determined, as well as the delay time relative to the same system clock of the actual path activated by the test pattern. Two metrics are then generated based on 1) the target path""s delay time, 2) the longest sensitizable path""s delay time, and 3) the application speed of the test on the IC tester with respect to the IC""s operating speed specification.
In accordance with the preferred embodiment, the metric for Delay Defect Exposure (DDE) is determined as follows. The slack times associated with (1) a longest sensitizable path through the IC that includes the delay fault site and (2) the actual path exercised by the test pattern through the IC that includes the delay fault site are determined. A difference between these slack times is then obtained. The difference is then summed with the difference between (3) the speed at which the test is performed and (4) the design specification operating speed of the IC for the actual path. The sum represents the DDE quality metric associated with the test for this delay fault. The DDE indicates the size (i.e., time duration) of a delay defect that could escape the delay test but still cause some other path on the IC to fail in normal operation.
Preferably, the DDE is calculated for each fault site using the set of test patterns associated with each fault site to enable each test to be xe2x80x9cgradedxe2x80x9d with respect to each delay fault. The resultant metrics for each fault site can be collected using standard statistical measures of mean, median, minimum, maximum, standard deviation, etc., to form IC-wide DDE metrics. The smaller the DDE value, the higher the quality of the test set at detecting small delay defects. In converse, the larger the DDE value, the lower the quality of the test set at detecting small delay defects.
In accordance with the preferred embodiment, the metric for Delay Sensitivity Ratio (DSR) is determined. To determine the DSR metric, the slack times associated with (1) the actual path exercised by the test pattern through the IC that includes the delay fault site is determined, and (2) a longest sensitizable path through the IC that includes the delay fault site is determined. The ratio between these slack times is then obtained. The ratio represents the DSR quality metric associated with the test for this delay fault. The DSR indicates the relative effectiveness of the actual path used to test the delay fault site with respect to the longest sensitizable path through that site.
Preferably, the DSR is calculated for each delay fault site using the set of test patterns associated with each fault site, thereby enabling each test to be xe2x80x9cgradedxe2x80x9d with respect to each delay fault. The resultant metrics for each fault site can be collected using standard statistical measures of mean, median, minimum, maximum, standard deviation, etc., to form IC-wide DSR metrics. The closer the DSR value is to 1, the higher the quality of the test set is at detecting small delay defects. In converse, the closer the DSR value is to 0, the lower the quality the test set is at detecting small delay defects.
Therefore, by taking these different timing factors into account, a true quality metric can be derived for each test pattern with respect to each delay fault. Furthermore, the entire set of tests can be graded by statistically combining the component quality metrics at each fault site together to obtain an overall quality metric for the test set. Additionally, for each fault site, the individual test having the lowest DDE (and correspondingly the highest DSR) will be the best test pattern for testing that delay fault site.
These and other features and advantages will become apparent from the following description, drawings and claims.